RAKE receivers receive various multipath signals through a plurality of relevant detectors and combine the signals together. In the Wideband Code Division Multiple Access (WCDMA) downlink receiving, due to the inter-symbol interference (ISI) and multi-user interference caused by multipath, the performance of the RAKE receivers are affected, especially at a high data transfer rate (for example, High Speed Downlink Packet Access, HSDPA), because the spreading factor is relatively smaller, not only the anti-interference capability of the RAKE receivers is reduced, but also the diversity gain introduced by multipath combination is greatly reduced. Therefore, in the environment of WCDMA high data transfer rate, in order to achieve better performance, a linear minimum mean square error (LMMSE) equalizer receiving technology is usually adopted. The principle of the LMMSE equalizer is to minimize the mean square error between an outputted equalization signal after being processed by the LMMSE equalizer and the transmitting signal, so as to eliminate the interference caused by multipath.
Hereinafter, the working manner of the LMMSE equalizer in the transmitting diversity mode is introduced.
In WCDMA, the transmitting diversity mode includes space-time coding transmitting diversity and closed-loop transmitting diversity. In the space-time coding transmitting diversity, double antennas are adopted for transmission, and the coding manner is shown in FIG. 1, in which Antenna 1 and Antenna 2 transmit symbols a1 and a2 respectively. The data transmitted by Antenna 1 is kept unchanged after passing through a space-time encoder, and the data transmitted by Antenna 2 is coded into −a2* and a1* by the space-time encoder and then transmitted.
FIG. 2 shows a modulation mode of the closed-loop transmitting diversity. A dedicated physical control channel (DPCCH) signal and a dedicated physical data channel (DPDCH) signal are spread and scrambled after passing through the dedicated physical channel (DPCH), and then, the spread and scrambled DPCCH signal and DPDCH signal are weighted by w1 and w2, and then transmitted together with a common pilot channel (CPICH1) signal and a CPICH2 signal respectively through two antennas. w1 is a scalar constant, and w2 is a variable complex number, which is adapted to adjust the amplitude and phase of the data transmitted by Antenna 2, so as to maximize the power of the signal received by the receiver. w2 is determined by a user equipment (UE) terminal, and then the information about w2 is fed back to the base station. The common pilot signals CPICH1 and CPICH2 are still coded and transmitted in the manner of space-time coding, as shown in FIG. 3.
As shown in FIG. 4, an information signal x(k) is filtered by a shaping filter (root raised cosine (RRC) filter) and then transmitted. After the signal passes through a multipath fading channel overlapped with additive white Gaussian noise (AWGN), the receiver performs a receiving RRC filtering on the receiving signals to get a signal y(k). A channel estimation module receives the common pilot signal among the receiving signals to get channel estimation value H, and an equalizer weight calculation module calculates two equalizer weights wd1 and wd2 according to the channel estimation value H. The LMMSE equalizer 1 and LMMSE equalizer 2 perform an LMMSE equalization process on the signal y(k) through using wd1 and wd2, so that the equalization signals passing through the two LMMSE equalizers meet the requirement of minimizing the mean square error between the equalization signals and the transmitting signals of the two transmitting antennas. A demodulation module performs a processing on the equalized signals.
The received signal may be expressed as:y(k)=Γ1x1(k)+Γ2x2(k)+n(k)  (1),
in which k represents a chip sequence number;y(k)=[y1(k), . . . yP(k), . . . , . . . ,y1(k+E−1), . . . ,yP(k+E−1)]T,
in which yP(k) represents the pth sample of the kth chip for the receiving signal;xi(k)=[xi(k),xi(k+1),xi(k+2) . . . , . . . ,xi(k+E+L−2)]T,
which represents the vector of the signal transmitted by the ith antenna;
Γi represents a channel array, and the subscript represents the channel where the signal of the ith antenna passes through; and
      Γ    i    =      [                                                      h                              1                ⁢                                                                  ⁢                i                                      ⁡                          (                              L                -                1                            )                                                …                                                    h                              1                ⁢                                                                  ⁢                i                                      ⁡                          (              0              )                                                                                                                                                            ⋮                                                                          ⋮                                                                                                                                                                h              Pi                        ⁡                          (                              L                -                1                            )                                                …                                                    h              Pi                        ⁡                          (              0              )                                                                                                                                                                                                                                                            ⋱                                                                                                                                                                                                                                                                h                              1                ⁢                                                                  ⁢                i                                      ⁡                          (                              L                -                1                            )                                                …                                                    h                              1                ⁢                                                                  ⁢                i                                      ⁡                          (              0              )                                                                                                                                                            ⋮                                                                          ⋮                                                                                                                                                                h              Pi                        ⁡                          (                              L                -                1                            )                                                …                                                    h              Pi                        ⁡                          (              0              )                                            ]  
in which hpi(l−1) (l=1, . . . ,L), (p=1, . . . ,P) is a channel coefficient of the pth sample of the lth chip on the ith antenna, and L is a channel delay spread in a unit of a chip.
The equalized signals and the transmitting signal on each of the two antennas respectively meet the minimum mean square error:wd1=arg minE{∥wd1y(k)−x1(k+d)∥2}  (2),wd2=arg minE{∥wd2y(k)−x2(k+d)∥2}  (3).
As derived from Weiner Optimization Formula, the following equation is obtained:wdi=E{xi(k+d)yH(k)}E{y(k)yH(k)}−1 i=1,2  (4),
in which i represents a corresponding antenna number.
Considering the equalization weight of Antenna 1, Formula (1) is substituted into Formula (4), and as x1(k) and n(k) are irrelevant to each other, the following two circumstances are considered respectively:E{x1(k+d)yH(k)}=E{x1(k+d)x1H(k)}+E{x1(k+d)x2H(k)}  (5)E{y(k)yH(k)}=Γ1E{x1(k)x1H(k)}Γ1H+Γ2E{x2H(k)x2H(k)}Γ2H+E{n(k)nH(k)}+Γ1E{x1(k)x2H(k)}Γ2H+Γ2E{x2(k)x1H(k)}Γ1H  (6).
As known from the coding manners of the space-time coding transmitting diversity and the closed-loop transmitting diversity, the transmitting signals of the two transmitting antennas are not completely independent and irrelevant. However, as the transmitting signals are combined by many users and code channels, in the closed-loop transmitting diversity mode, each user has a different weight, and the transmitting diversity adopted by each code channel is not the same, so it is difficult to get the relevant value between the transmitting signals of the two antennas, and thus approximation is introduced, in which the signals of the two transmitting antennas are assumed to be independent and irrelevant, that is,
E{x1(k+d)x2(k)}=0, and d is an arbitrary number.
Under such approximation condition,E{x1(k+d)yH(k)}=edΓ1Hσx12  (7),E{y(k)yH(k)}=Γ1Γ1Hσx12+Γ2Γ2Hσx22+σn2  (8).
In the transmitting diversity mode, the transmitting powers of the two antennas are the same, which is σx2, so that:wd1=edΓ1H{Γ1Γ1H+Γ2Γ2H+σn2/σx2}−1  (9).
Similarly, as for transmitting Antenna 2,wd2=edΓ2H{Γ1Γ1H+Γ2Γ2H+σn2/σx2}−1  (10).
An approximation assumption is introduced into the above deriving process, in other words, regarding the signals of the two transmitting antennas as being completely irrelevant. However, in practice, the approximation cannot be ignored. If the last two items in Formula (6) are ignored, it inevitably causes loss of the performance. As can be seen from Formulae (9) and (10), the expressions for the taps of the two equalizers are quite similar to each other. As the transmitting antennas are generally very close to each other, the channel of Antenna 1 and the channel of Antenna 2 are similar to each other. In this case, the equalization signals output from the two equalizers inevitably contain the signals of the two antennas.
As the mean square error between the output of the equalizer 1 and the transmitting signal of the transmitting antenna 1 is at the minimum level and that between the output of the equalizer 2 and the transmitting signal of the transmitting antenna 2 is also at the minimum level, as shown in FIG. 5, the two equalization signals are equivalently considered as two transmitting signals, and after being descrambled and dispread, they are respectively decoded and demodulated according to the coding manner of the transmitting signals. The CPICH signal adopts the space-time coding manner in the two transmitting diversity modes, so the CPICH signal is decoded and demodulated in the space-time coding transmitting diversity mode under the two transmitting diversity modes. Therefore, for Antenna 1, the channel estimation value is obtained by directly being divided by a pilot symbol; for Antenna 2, the channel estimation value is obtained by being divided by a pilot symbol after odd-even reversal.
The high-speed physical downlink shared channel (HS-PDSCH) signal after being descrambled and dispread is approximately considered as a pattern merely containing one antenna. For the space-time coding transmitting diversity mode, the equalized signal conversion is shown in FIG. 6. The noises of the two transmitting antennas are considered as approximately the same, and the signals are multiplied by weight factors according to the signal-to-noise ratio, and then the same transmitting symbols are combined. The signal of Antenna 1 at the time t+1 and the signal of Antenna 2 at the time t are combined, and the signal of Antenna 2 at the time t+1 and the signal of Antenna 1 at the time t are combined. In the closed-loop transmitting diversity mode, because the modulation patterns are different, as shown in FIG. 7, the signals of Antenna 1 and Antenna 2 at the corresponding time are combined.
However, the two equalization signals are considered as signals respectively decoded according to one antenna, and the transmitting signals of the other antenna are considered as noise signals. Because the equalizers cannot completely convert the signals of the other antenna into random noises, the actual equalization signals are still blended with signals of the other antenna, which may results in loss of information. Furthermore, the combination manner changes the form of receiving signals, which causes that the diversity gain effect of the closed-loop transmitting diversity mode cannot be fully exerted.
Another solution, as shown in FIG. 8, in which an equal gain combination is directly performed after equalization, and then, after the combination, the channel estimation value and the demodulated signal are obtained in the conventional closed-loop transmitting diversity mode or the space-time coding transmitting diversity mode, and then a soft value is obtained and provided to the decoding module. In such a technical solution, the blending of the equalized signals from the two antennas is taken into consideration, instead of simply considering the equalized signals as signals from one transmitting antenna in the solution above. However, as the signal-to-noise ratios of the two antennas are different, if the equal gain combination is directly performed on the equalized signals, the maximization of signal-to-noise ratio is not realized, and when the noise powers of the two signals are different from each other, the performance of the demodulated signal is not ideal.
It can be seen from the above description that, the performances of the demodulated signal obtained through the two solutions are not ideal.